Stripline Impedance Calculator
A trace buried between two reference planes. Unlike microstrip, the field is entirely inside the dielectric, so the effective Dk is simply the substrate Dk and the line is genuinely non-dispersive. This page evaluates the exact zero-thickness conformal-mapping result rather than a curve fit.
Choosing something to solve for makes Characteristic impedance an editable target and computes the chosen input from it.
Formula
| w | trace width (m) |
| b | plane-to-plane spacing (m) |
| er | substrate relative permittivity |
| K | complete elliptic integral of the first kind |
| k, k' | conformal-mapping modulus and its complement |
What this model assumes, and where it stops
Assumptions
- Quasi-TEM propagation: the cross-section is small against a wavelength.
- Static (low-frequency) solution - no dispersion, so Dk and Z0 do not change with frequency.
- Perfect conductors and a lossless, homogeneous, isotropic dielectric.
- Infinitely wide reference plane with no splits, voids or stitching gaps under the trace.
- A uniform, straight, infinitely long line - no bends, vias, stubs or terminations.
- The trace is exactly centred between the two planes.
- Zero conductor thickness - this is the mathematically exact limit.
- Homogeneous dielectric above and below with the same Dk.
Limitations
- Zero-thickness only. Real copper raises the capacitance and lowers Z0; at 35 um copper in a 0.5 mm stack that is worth several percent, and this page does not correct for it.
- An off-centre trace (asymmetric stripline) has a different impedance - the further off centre, the lower the Z0.
- Prepreg and core often have different Dk. This model takes one value for the whole gap.
- No conductor or dielectric loss, so no attenuation output.
- Glass-weave skew is invisible here: the model sees a homogeneous dielectric, a real laminate does not.
When you need a 3D field solution instead
Closed-form models like the one above hold on idealised geometry. These are the cases where they stop being good enough and a full 3D electromagnetic and thermal solution is the only way to get a trustworthy answer:
- Reference-plane splits, voids or antipads under the trace, which the infinite-plane assumption cannot see.
- Vias, layer transitions and connector launches, where the discontinuity dominates the impedance budget.
- Tight bends, taper transitions and length-matching serpentines.
- Anything above roughly 5 GHz, where dispersion, surface roughness and dielectric loss stop being negligible.
- Trace-to-trace coupling beyond a single adjacent pair, and coupling between layers.
- Etch-factor trapezoidal cross-sections, where the top and bottom widths differ enough to shift Z0 by several percent.
- Asymmetric or dual stripline stacks, where the two planes are at unequal distances.
- Mixed-dielectric stacks where core and prepreg Dk differ enough to matter.
Common questions
Why is stripline slower than microstrip on the same laminate?
Because the field is entirely inside the dielectric. Microstrip has part of its field in air, so its effective Dk is lower and the wave travels faster. On FR-4 that is roughly 6.7 ns per metre for stripline against 6.0 for microstrip.
What width gives 50 ohm stripline?
Around 0.47 times the plane-to-plane spacing on Dk 4.3, so about 0.76 mm in a 1.6 mm gap. The ratio is scale-invariant, so halving the stack halves the width.
Does this handle asymmetric stripline?
No. This is the symmetric case with the trace exactly centred. Moving the trace off centre lowers the impedance, and the further off centre it sits the larger the error in using this result.
References
- Cohn - Characteristic Impedance of the Shielded-Strip Transmission Line, IRE MTT 1954
- Wadell - Transmission Line Design Handbook, Artech House 1991